 |
|
| If you can't view the Datasheet, Please click here to try to view without PDF Reader . |
|
|
|
| If you can't view the Datasheet, Please click here to try to view without PDF Reader . |
|
|
| Datasheet File OCR Text: |
| november 2010 doc id 17975 rev 1 1/22 AN3276 application note st solution for efficiency im provement in pfc applications, back current circuit (bc 2 ) introduction the challenges for modern high efficiency switching power supplies are to minimize power losses and increase their power density without raising the cost. the goal is to reduce both power conduction and power switching losses. minimization of power conduction losses is difficult to achieve without considerably affecting the cost and power density, since more material is required (bigger active and passive components). unlike the conduction losses, it is easier to reduce the power switching losses without significantly increasing the power supply cost. there are two main ways to achieve this improvement: working on the dynamic behavior of the semiconductor technologies working on circuit topologies novel diodes using technologies such as sic and gan materials significantly reduce the switching losses. however, their high price makes them not so attractive for applications such as desktop server power supplies, solar inverters and inverters. the patented circuit [see section 5: references , 1. ], described in this application note is based on the soft switching method and meets market expectations since its efficiency/cost/power, and density/emi trade-offs are better than high voltage sic schottky diodes. www.st.com
contents AN3276 2/22 doc id 17975 rev 1 contents 1 existing solutions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 1.1 diode switch-on losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 1.2 soft switch-on method . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 1.3 active recovery circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 1.4 passive recovery circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 2 the new st solution - bc 2 : energy recovery circuit . . . . . . . . . . . . . . . 7 2.1 concept description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 2.2 phase timing description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 2.2.1 phase before t0 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 2.2.2 phase t0 to t1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 2.2.3 phase t1 to t2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 2.2.4 phase t2 to t3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 2.2.5 phase t3 to t4 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 2.2.6 phase t 4 to t 5 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 2.3 electrical voltage stress in bc 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 2.4 calculation of m2 and m1 ratios . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 2.5 calculation of l . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 2.6 range of products . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 3bc 2 design in 450 w pfc . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 3.1 bc 2 design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 3.2 bc 2 typical waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 3.3 efficiency comparison . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 3.4 thermal measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 4 conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 5 references . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 6 revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 AN3276 existing solutions doc id 17975 rev 1 3/22 1 existing solutions this section describes some existing areas for efficiency improvements in pfc applications. 1.1 diode switch-on losses usually, in mass market applications between 200 w and 2 kw, a power factor corrector (pfc) working in continuous conduction mode (ccm) is mandatory. to improve the power converter density, the switching frequency should be increased. nevertheless, when the switching frequency increases, power dissipation in the power switch/rectifier commutation cells leads to the major switching losses in the pfc. the main power losses occur during turn-on of the power switch due to both the voltage and current crossing area of the mosfet and the reverse recovery losses [see section 5: references , 2. ] produced by the pn diode as shown in figure 1 . figure 1. switch-on losses in pn diode behavior to reduce the losses of the pn rectifier, many semiconductor manufacturers have recently introduced high-voltage schottky diodes using sic and gan technologies. however, it is impossible to completely remove the voltage and current crossing area during transistor turn-on by improving component performance only. t losses are due to area between i +i and v 0rm ds i+i rm 0 i 0 v ds w on existing solutions AN3276 4/22 doc id 17975 rev 1 figure 2. switch-on losses in sic or gan diode behavior unlike pn diodes, sic diodes allow the turn-on di/dt to be increased without increasing the diode recovery current. thus, switching time decreases and switch-on losses decrease too, but they are not removed entirely. today, in pfc designs, the turn-on di/dt with the sic diode is around 1000 a/s maximum to respect emi st andards, whereas the pn diode is used with a di/dt of 300 a/s. 1.2 soft switch-on method another way to reduce these losses is to use a soft switching method by adding a small inductor l to control the di/dt slope. this solution removes the current/voltage crossing area and the pn diode recovery current effect during the turn-on of the transistor as shown in figure 3 . figure 3. switch-on losses in current soft switching behavior t i 0 v ds losses are due to area between i and v 0ds t small losses: zero current switching i+i rm 0 i 0 v ds di/dt fixed by l AN3276 existing solutions doc id 17975 rev 1 5/22 this soft switching solution is well known, but it requires that several technical criteria be met: reset the current in the inductor l at each switching period, whatever the variations of the current, and input and output voltages. recover the saved inductive energy without losses. limit any overvoltage and overcurrent stress in the semiconductor devices. keep cost down when adding any device. maintain a similar po wer supply density. there are many circuits that are classifi ed in two families of recovery circuits: active passive 1.3 active recovery circuit in the active recovery circuit family, the zero voltage transition (zvt) [see section 5: references , 3. ] shown in figure 4 is well known by designers. this circuit allows both switch-on and switch-off powe r losses to be removed. figure 4. zero voltage transition (zvt) active recovery circuit a theoretical study indicates that zvt is an ex cellent topology for the pfc application, since all the switch losses are removed. in addition, this circuit can work whatever the input and output power variations. nevertheless, in practice, the recovery current from the boost diode d b significantly affects the zvt behavior leading to some constraints on both inductance and minimum duty cycle. during the reset current in the small inductor l, the recovery current from d 2 involves a high-stress voltage and damp ing parasitic oscillation. finally, the dynamic behavior of the pn diode affects th e global zvt efficiency because conduction times in the transistor should increase and a dissipative snubber is mandatory to reduce the electrical stress across the semiconductors. in terms of cost the zvt circuit requires an additional power mosfet and a specific pwm controller. several derivative circuits of the zvt circuit have the same technical issue and their higher price makes these circuits less than ideal for mass market applications. therefore, the passive recovery circuit can be more attractive. v out v mains d b d 2 d 1 l t zvt t r l b r damping d damping existing solutions AN3276 6/22 doc id 17975 rev 1 1.4 passive recovery circuit in the passive recovery circuit family the electrical schematic shown in figure 5 is a good example [see section 5: references , 4. ]; only two extra diodes and one resonant capacitor are required. figure 5. passive recovery circuit this circuit works well under unchanging external conditions. however, it is difficult to design this kind of system in pfc applications since the current reset in the small inductor depends on both boost diode recovery current and the external electrical conditions. although, the non-dissipative passive circuit requires fewer components, it is unfortunately technically impractical in pfc applications. these examples highlight that the current snubber method is well known but the technical challenge is to recover the l energy through the application without affectin g the five criteria listed in section 1.2 . v mains v out d 2 d 1 d b l l b t r c r AN3276 the new st solution - bc 2 : energy recovery circuit doc id 17975 rev 1 7/22 2 the new st solution - bc 2 : energy recovery circuit the innovative circuit has been designed [see section 5: references , 1. ] to respect the five soft switching criteria in section 1.2 . figure 6 shows that two additional diodes d 1 and d 2 and two auxiliary windings n s1 and n s2 wound around the main boost inductor l b are designed to reset the energy stored in the small inductor l. figure 6. novel energy recovery circuit: bc 2 2.1 concept description the winding n s1 allows the i rm current from the boost diode d b to be recovered in the main boost inductor when the transistor turns on. since the mains input voltage modulates the l b voltage, it also modulates the reflected voltage across n s1 . this input voltage also modulates the boost diode current i db and its associated recovery current i rm . these combined modulations allow the extra current i rm flowing in the inductor l to be reset into the winding n s1 even in the worst case. the winding n s2 allows the extra current of l to be injected into the output capacitor when the transistor turns off. the reflected voltage across n s2 is also a function of the input voltage. this reflected voltage reaches its maximum when the ac line voltage is low, corresponding to the maximum value of the inductor l current. these combined variations allow the current flowing in the inductor l to be cancelled in the bulk capacitor through the diode d 2 even in the worst case. the benefits of these two additional windings n s1 and n s2 are to switch off the diodes d 1 and d 2 with a low di/dt (about 10 a/s) as in a discontinuous mode switching converter. their recovery currents do not affect the behavior of the bc 2 circuit. l n s2 m 2 =n s2 /n p d b d 1 t r m 1 =n s1 /n p d 2 n s1 n p l b v out v mains the new st solution - bc 2 : energy recovery circuit AN3276 8/22 doc id 17975 rev 1 2.2 phase timing description figure 7. equivalent timing per phase the winding ratios m 1 and m 2 versus n p winding are those of the windings n s1 and n s2 respectively. l ) m ( v m v dt di mains out tr + + - = 2 2 1 2 2 1 + m m v v main out - ) m 1 ( m v 1 1 mains - - ) m 1 ( m v v 2 2 mains out + + 2 2 out mains m m 1 ) v v ( + - ) m 1 ( ) m 1 ( i 1 2 rm - + ? ? ? ? ? ? - + + - 1 2 1 mains out m 1 m m v v l ) m 1 ( v m v dt di 2 2 mains 2 out db + + - = 1 2 1 1 2 1 2 m m m m m i rm - - - v 0ut i 0 i 0 i 0 i 1 i 1 i 1 -i rm (i rm +i 0 )(1+m 2 ) t d 1 t d 2 i 0 i tr v tr v tr v l v l v db v db i d2 i d1 i db i mains t t t t t t t t 5 t 4 t 3 t 2 t 1 t 0 AN3276 the new st solution - bc 2 : energy recovery circuit doc id 17975 rev 1 9/22 2.2.1 phase before t 0 figure 8. equivalent circuit before t 0 before t 0 , the bc 2 circuit has the same behavior as the conventional boost converter. the boost diode d b conducts to send the main inductor energy through the output bulk capacitor. 2.2.2 phase t 0 to t 1 figure 9. equivalent circuit t 0 to t 1 at t 0 , the power mosfet turns on and the current in d b is equal to i 0 . at t 0 +, the current soft switching occurs, that is to say, the voltage across the power mosfet decreases to 0 volt under a zero current and no switching losses appear. after t 0 , the current flowing in l increases linearly until it reaches the input current i 0 added with the recovery diode i rm , whereas the current flowing in d b decreases linearly down to -i rm . figure 7 shows the behavior of these currents taking account of the m 2 transformer ratio. the simplified di/dt expression in transistor t r and the boost diode d b can be estimated using , and at t 0 +, the c oss capacitance of the power mosfet is discharged in its r ds(on) . unlike standard pfc circuits, the voltage applied across the drain is lower because the reflected c oss l d b d 2 d 1 v mains n s2 n s 1 i 0 v out + - - + c oss l d b d 2 d 1 v v mains n s2 n s 1 i i i 0 0 rm v out t r + - - + l v v dt di dt di n s2 out tr db ? = 2 m 1 2 m ). v v ( v mains out 2 n s + ? = the new st solution - bc 2 : energy recovery circuit AN3276 10/22 doc id 17975 rev 1 v ns2 voltage is subtracted from v out . this behavior provides the bc 2 circuit with a benefit since under the low output load, power saving occurs in the system and it can be evaluated using: thus, the bc 2 reduces switch-off losses too. 2.2.3 phase t 1 to t 2 figure 10. equivalent circuit t 1 to t2 at t 1 +, the boost diode d b turns off, and an overcurrent i rm is stored in the small inductor. this overcurrent discharges the d b junction capacitance linearly. at the same time, the voltage polarity across the main inductor changes until it reaches d 1 diode conduction. at this time, the overcurrent i rm is reduced by the transformer ratio m 1 and is sent to the main inductor. thus, the current flowing through n s1 contributes to charge the internal coil magnetization l b at the same time as the n p winding biased by the mains voltage. the i rm current flowing in d 1 decreases down to reach 0 a thanks to the reflected voltage v ns1 that it is given by: and to guarantee a soft switching behavior in discontinuous mode, the current in d 1 should reach 0 a before time t 3 . the t d1_on time trend supports the pfc application since the i rm current is the largest when the v mains voltage in the sinusoidal period is the highest. in addition, to cancel the d 1 recovery current diode effect, the di/dt_ d1 is always low thanks to the low reflected voltage v ns1 and it is given by: unfortunately, during this phase a high reverse voltage is applied across the boost diode d b : switching 2 mains out 2 out oss 0 t oss c f . 2 m 1 2 m . v v v . c . 2 1 p ? ? ? ? ? ? ? ? ? ? ? ? ? ? + + ? = ? l d b d 2 d 1 v v mains n s2 n s 1 i i 0 rm v out t r + - - + c j 1 m 1 1 m . v v mains 1 ns ? ? = 1 m . v ) 1 m 1 .( l . i _ t mains rm on 1 d ? l ). 1 m 1 ( 1 m . v dt di _ d mains ? ? 1 1 m 1 ) 2 m 1 m .( v v v v v v mains out 2 n s 1 n s out reverse _ db ? + + = + + = AN3276 the new st solution - bc 2 : energy recovery circuit doc id 17975 rev 1 11/22 this feature requires a specific diode for th is application, so st has developed an optimized diode with an accurate trade-off between the i rm current value and its breakdown voltage. 2.2.4 phase t 2 to t 3 figure 11. equivalent circuit t 2 to t 3 at t 2 , the current in d 1 reaches 0 a and then the bc 2 works as a conventional power boost converter. as the power transistor stays on, the current in the main l b and the small l inductor increases up to i 1 at time t 3 . 2.2.5 phase t 3 to t 4 figure 12. equivalent circuit t 3 to t 4 at t 3 , the power transistor turns off. at this time the voltage across the mosfet increases linearly as its c oss capacitance is charged by the current stored in the small inductor l to reach conduction in diode d 2 . no overvoltage stress occurs on the power switch during the turn off. at the same time, the voltage polarity across the main inductor changes until it reaches d b diode conduction. as soon as the diodes conduct together, the output current is shared as shown in figure 7 . the current in d 2 starting at i 1 , decreases due to the reflected voltage from n s2 to reach 0 a with a low di/dt. on the other hand, the current in d b rises to reach the nominal current at t 4 . this shared current is a benefit for the bc 2 circuit. in the pfc application working under a lower mains voltage such as 90 v rms, the highest l d b d 2 d 1 v v mains n s2 n s 1 i 1 v out t r + - - + c j c oss l d b d 2 d 1 v v mains n s2 n s 1 i 1 v out + - - + the new st solution - bc 2 : energy recovery circuit AN3276 12/22 doc id 17975 rev 1 application boost current is shared between d b and d 1 diodes. therefore the conduction losses in the rectification stage are reduced. the reflected voltage v ns2 and d 2 time conduction are given by: and the t d2_on time trend supports the pfc application since the i 1 current is largest when the v mains voltage range is lowest. thus the discontinuous mode could be guaranteed in the bc 2 circuit even under the worst case pfc applications such as high output load current under the minimum v mains voltage range. in addition, to cancel the d 2 recovery current diode effect, the di/dt_d 2 is always low due to the low reflected voltage v ns2 given by: 2.2.6 phase t 4 to t 5 figure 13. equivalent circuit t 4 to t 5 at t 4 , the current in the d 2 reaches 0 a and then the bc 2 works like a conventional power boost converter. only the boost diode d b conducts. due to the reflected voltage of n s2 , the voltage across the power switch is lower than v out . thus the c oss capacitor is discharged in the bulk capacitor and power saving oc curs as the transistor turns on at t 0 . 2 m 1 2 m ). v v ( v mains out 2 n s + ? ? = 2 m ). v v ( ) 2 m 1 .( l . i t mains out 1 on _ 2 d ? + ? = l ). 2 m 1 ( 2 m ). v v ( dt d _ di mains out 2 + ? ? c oss l d b d 2 d 1 v mains n s2 n s 1 i 0 v out + - - + AN3276 the new st solution - bc 2 : energy recovery circuit doc id 17975 rev 1 13/22 2.3 electrical voltage stress in bc 2 ta bl e 1 summarizes the maximum voltage across each semiconductor versus the phases. the bc 2 circuit needs to use a specific diode with a breakdown voltage higher than 600 v. moreover, its recovery current should be optimized to avoid an higher current in the power transistor during the phase [t 1 -t 2 ]. st has developed specific diodes (stth16bc065c, stth10bc065c and stth8bc065) for the bc 2 circuit. they have been designed to sustain average currents of 5 to 8 a (depending on the application) and a repetitive reverse voltage v rrm of 650 v. 2.4 calculation of m 2 and m 1 ratios to respect the discontinuous operating mode during the timing phases [t 1 -t 2 ] and [t 3 -t 4 ], the time td1 and td2 shown in figure 7 should be always positive. according to the typical ccm pfc rules and both t d1_on and t d2_on expressions, it becomes easy to define the m 1 and m 2 transformer ratio conditions. and where p in is the input pfc power, f s is the switching frequency, v mainsrms max is the maximum rms voltage range and the i rm max is the maximum current recovery under the turn-on di/dt at its maximum operating junction condition. table 1. maximum reverse voltage in the bc 2 maximum voltages phases waveform expression > v out ? vr trmax t 3 - t 4 v out no vr dbmax t 1 - t 2 yes vr d1max t 0 - t 1 t 3 - t 4 no vr d2max t 0 - t 1 t 1 - t 2 t 2 - t 3 v out no out 1 2 1 max mainsrms v m 1 ) m (m 2 v + + + ? ? out 2 2 1 out max mainsrms v m 1 ) m (m ) v 2 (v + + + ? ? ? () () l v max p 2 max v - v max v f l v max p m out in mainsrms out 2 mainsrms s out in 2 ? ? ? ? ? ? ? > ? () () 2 max v v 2 max v f v l max i m 1 m mains out mains s out rm 2 1 ? ? ? ? ? ? ? ? + > the new st solution - bc 2 : energy recovery circuit AN3276 14/22 doc id 17975 rev 1 2.5 calculation of l there are several ways to rate the inductance l. for instance, its turn-on di/dt can be rated at 50 a/s. then, m 2 and m 1 are calculated taking into account the corresponding i rm of diode d b . however, the reverse voltage across the d b, v rdb _reverse, must not exceed 75% of v rrm to meet the system design rule, 75% x 650 = 487 v. if v rdb _ reverse is higher than 487 v, the l value should be reduced. therefore, the di/dt of l and the i rm diode of d b increase as well. thus, m 1 and m 2 should be recalculated to get v rdb_reverse below 487 v. but this calculation method does not optimize the inductance l and its size. ultimately, a good rating should minimize l size. st has developed a software tool using all the parameters: the i rm diode of d b versus the di/dt and junction t j , the l inductance tolerances, and the switch-on power losses. this tool is proposed to help designers to choose the best l inductance for its application. ta b l e 2 shows two pfc examples using the bc 2 concept. 2.6 range of products st offers the bc 2 technology in a range of products: stth8bc065di, stth8bc060d, stth5bcf060 for applications from 800 w to 2 kw stth16bc065ct, stth5bcf060 for applications from 400 w to 1 kw stth10bc065ct + stth3bcf060u for applications from 280 w to 600 w table 2. inductor and size versus the pfc types case 1 case 2 input power 100 w < pin < 1.1 kw 50 w < pin < 450 w v mains rms 90v < v mains rms < 264 v 90 v bc 2 design in 450 w pfc AN3276 16/22 doc id 17975 rev 1 3.3 efficiency comparison bc 2 and sic diode efficiency have been compared under two v mains levels as shown in figure 15 (230 v rms ) and figure 16 (90 v rms ) with a switching frequency equal to 140khz. at 230v rms , the bc 2 circuit saves up to 2.25 w at full load and 1 w at 100 w compared to the 8 a sic diode. under low load, the reflected voltage from n s2 still improves the bc 2 efficiency because the switch-off losses are lower than the sic as described in the phase time [t 0 -t 1 ]. as soon as the pfc works in disc ontinuous mode (< 100 w) the sic and the bc 2 have the same efficiency as shown in figure 15 . figure 15. efficiency comparison at 230 v rms at 90 v rms, the soft switching method benefits plus the power saving in the c oss discharge reinforces the benefit of the bc 2 circuit. up to 5.4 w is saved thanks to the bc 2 at 450 w compared to the sic diode and under low load, up to 1.7% is saved thanks to the switch-off power saving. 88 89 90 91 92 93 94 95 96 97 50 100 150 200 250 300 350 400 450 p out % bc 2 solution 8a sic continuous mode discontinuous mode sic bc2 +1% +0.7% +0.5% fs = 140 khz efficiency = f(p ) out v = 230 v mains rms AN3276 bc 2 design in 450 w pfc doc id 17975 rev 1 17/22 figure 16. efficiency comparison at 90 v rms figure 17. 450 w pfc efficiency versus three different output powers and three switching frequencies for v mains rms = 90 v figure 17 highlights the benefit of the bc 2 circuit soft switching method added to its c oss discharge power saving especially at low load. 83 84 85 86 87 88 89 90 91 92 93 94 95 50 100 150 200 250 300 350 400 450 % bc2 solution 8a sic continuous mode sic bc2 +1.7% +1.5% +1.2% fs = 140 khz v = 90 v mains rms efficiency = f(p ) out p out efficiency=f(pout,fs) 80 81 82 83 84 85 86 87 88 89 90 91 92 93 94 95 52 203 400 pout % bc2 solution 72khz 8a sic 72khz bc2 solution 144khz 8a sic 144khz bc2 solution 200khz 8a sic 200khz 72khz 72khz 72khz 140khz 140khz 200khz 200khz 200khz 140khz +0.7% +1.7% +2% p out =50w p out =200w +0.5% +1.5% +1.6% +0.4% +1.2% +1.5% efficiency=f(p out ,fs) p out =400w vmains=90vrms f s = f s = f s = bc 2 design in 450 w pfc AN3276 18/22 doc id 17975 rev 1 3.4 thermal measurement the soft current switching method allows the power dissipation in the switch transistor to be reduced. figure 18 shows that a high thermal temperature difference (18 c) occurs between the bc 2 and the sic diode in the pfc application. for the same working junction temperature in the power transistor, the size of the thermal heatsink could be reduced. in this way, the space saving balances out the space from the small l inductor required by the bc 2 circuit. therefore, the bc 2 circuit can have the same po wer density as the sic diode solution. nevertheless, the bc 2 efficiency decreases due to the power mosfet r ds(on) thermal rising. figure 18 shows that 0.75 w should be removed from the 5.4 w measured in the efficiency comparison at 90 v rms. even in this configuration, the bc 2 circuit has a better efficiency than the sic diode. the bc 2 circuit has a better power density and efficiency criteria than the sic diode. another consideration is that, for the same thermal heatsink, the power mosfet size could be reduced to further reduce the pfc cost. figure 18. thermal measurement comparison stp20nm50 stp20nm50 stth8bc065di tr db stth8bc065 diode in the bc2 circuit power mos thermal junction measurement bc2 circuit tj(avg)= 72 c rdson stp20nm50 rdson= 0.26 tr cond. losses (0.26/2)x4.282 =2.38w power mos thermal junction measurement sic diode tj(avg)= 90 c rdson stp20nm50 rdson= 0.34 tr cond. losses (0.34/2)x4.282 = 3.12w +0.75w stp20nm50 stp20nm50 stpsc806d tr db stpsc806d diode in the boost circuit pfc : v = 90 v mains rms , fs = 140 khz , p = 450 w , out i = 4.28 a , rg = 47 trrms 1 AN3276 conclusion doc id 17975 rev 1 19/22 4 conclusion the bc 2 circuit uses the soft switching method with a unique non-dissipative recovery circuit. stmicroelectronics has introduced the specific diodes suited for the bc 2 concept to improve the ccm pfc performance as shown in ta b l e 4 . this circuit supports the actual energy efficiency recommendation at 20%, 50% and 100% of rated power supplies. table 4. bc 2 benefits in 450 w pfc, 140 khz criteria at 90 v rms sic diode bc 2 circuit benefits efficiency 92.5% 93.7% +1.2% power mosfet temperature 90 c 72 c -18 c power density at iso-efficiency state of art improve, save 5.4 w +1.2% gain emi acceptable with di/dt < 1000 a/s acceptable with di/dt < 100 a/s soft switching boost rectification cost > 0.2 ? per 100 w >> 0.2 ? per 100 w mass market references AN3276 20/22 doc id 17975 rev 1 5 references 1. beno?t peron, ?auxiliary switching circuit for a chopping converter?, patent no: us 6,987,379 b2, june 2006 2. bertrand rivet, ?new solution to optimize diode recovery in pfc boost converter?, pcim 2000. 3. jim noon, uc3855a/b high performance power factor preregulator -texas instrument- application report- slua146a 4. brian t, irving and m. jovanovic ?analysis, design and performance evaluation of flying-capacitor passive lossless snubbe r applied to pfc boost converter?, apec 2002, pp. 503 - 508 vol.1. AN3276 revision history doc id 17975 rev 1 21/22 6 revision history table 5. document revision history date revision changes 10-nov-2010 1 initial release. AN3276 22/22 doc id 17975 rev 1 please read carefully: information in this document is provided solely in connection with st products. stmicroelectronics nv and its subsidiaries (?st ?) reserve the right to make changes, corrections, modifications or improvements, to this document, and the products and services described he rein at any time, without notice. all st products are sold pursuant to st?s terms and conditions of sale. purchasers are solely responsible for the choice, selection and use of the st products and services described herein, and st as sumes no liability whatsoever relating to the choice, selection or use of the st products and services described herein. no license, express or implied, by estoppel or otherwise, to any intellectual property rights is granted under this document. i f any part of this document refers to any third party products or services it shall not be deemed a license grant by st for the use of such third party products or services, or any intellectual property contained therein or considered as a warranty covering the use in any manner whatsoev er of such third party products or services or any intellectual property contained therein. unless otherwise set forth in st?s terms and conditions of sale st disclaims any express or implied warranty with respect to the use and/or sale of st products including without limitation implied warranties of merchantability, fitness for a parti cular purpose (and their equivalents under the laws of any jurisdiction), or infringement of any patent, copyright or other intellectual property right. unless expressly approved in writing by an authorized st representative, st products are not recommended, authorized or warranted for use in milita ry, air craft, space, life saving, or life sustaining applications, nor in products or systems where failure or malfunction may result in personal injury, death, or severe property or environmental damage. st products which are not specified as "automotive grade" may only be used in automotive applications at user?s own risk. resale of st products with provisions different from the statements and/or technical features set forth in this document shall immediately void any warranty granted by st for the st product or service described herein and shall not create or extend in any manner whatsoev er, any liability of st. st and the st logo are trademarks or registered trademarks of st in various countries. information in this document supersedes and replaces all information previously supplied. the st logo is a registered trademark of stmicroelectronics. all other names are the property of their respective owners. ? 2010 stmicroelectronics - all rights reserved stmicroelectronics group of companies australia - belgium - brazil - canada - china - czech republic - finland - france - germany - hong kong - india - israel - ital y - japan - malaysia - malta - morocco - philippines - singapore - spain - sweden - switzerland - united kingdom - united states of america www.st.com |
Price & Availability of AN3276
 |
|
|
|
|
All Rights Reserved © IC-ON-LINE 2003 - 2022 |
| [Add Bookmark] [Contact Us] [Link exchange] [Privacy policy] |
|
Mirror Sites : [www.datasheet.hk]
[www.maxim4u.com] [www.ic-on-line.cn]
[www.ic-on-line.com] [www.ic-on-line.net]
[www.alldatasheet.com.cn]
[www.gdcy.com]
[www.gdcy.net] |